Joint estimation of communication channel effects in communication receivers

ABSTRACT

A receiver signal is sampled at a sampling rate equivalent to a chip rate at which chips of a known signal are timed. The resulting receiver signal samples are segmented into receiver signal segments, which are filtered by respective matched filters that are matched to known signal segments segmented from the known signal. Indexes are assigned to elements of the resulting filter response sequences to define an array thereof. Frequency transforms are computed of elements of the filter response sequences in respective columns of the array. Indexes are assigned to elements of the resulting frequency response sequences to define another array thereof. Channel effects imparted on a radio signal are jointly estimated from characteristics of the other array at which at least one local maximum is located.

RELATED APPLICATION DATA

This application claims benefit of priority under 35 U.S.C. § 119(e)from U.S. Provisional Patent Application No. 62/780,841 entitled “JointAcquisition of Doppler and Delay for Direct Sequence Spread SpectrumSignals,” filed on Dec. 17, 2018, the entire disclosure of whichincorporated herein by reference.

BACKGROUND

Direct-sequence spread spectrum (DSSS) is a modulation technique used toreduce overall signal interference. With DSSS, the message signalcomprises message bits that are modulated by a bit sequence known as aPseudo Noise (PN) code (a pseudorandom sequence of −1 and 1 values). ThePN code is comprised of radio pulses commonly referred to as chips thatare much shorter in duration than the message bits. Such modulation ofthe message signal results in a signal that resembles white noise with abandwidth nearly identical to that of the PN sequence. The smaller thechip duration, the larger the bandwidth of the resulting DSSS signal,which results in better resistance against interference. The originalmessage bits may be reconstructed from this noise-like signal at areceiver by multiplying the signal by the same PN code in a processknown as “de-spreading,” where the transmitted PN sequence is correlatedwith the PN sequence known to the receiver.

Among the advantages of DSSS is its security; information conveyed usingDSSS techniques can be difficult to intercept without the PN code usedby the transmitter. But knowing the PN code alone may be insufficient torecover the message bits from the received signal in certain situations.Consider communication between a maritime/terrestrial mobile platform onthe open water/ground and a friendly aerial platform (e.g., aircraft,UAV, hypersonic vehicle and missile) flying quickly at a very low angleof elevation in order to evade enemy detection. A covert and reliablelow probability detection (LPD) communication link is desired withfriendly aircraft. The covert aspect comes from DSSS and other lowprobability detection waveform characteristics, while the reliabilityaspect comes from the receiver design and signal processing. A DSSSsignal can be despread at the receiver as long as the locally generatedcopy of the spreading signal is subject to the same delays, carrierfrequency offsets, and other imperfections (referred to herein ascommunication channel effects) that are seen by the transmitted signalas it is conveyed over a communication channel. Even small deviations indelay or frequency can prevent reliable despreading. The process ofmatching the frequency offset and delay of the spreading signal to thoseof the received signal is generally known as the “code synchronization”problem.

SUMMARY

To estimate channel effects on a radio signal conveyed over acommunication channel, sequential receiver signal samples are generatedby sampling the received signal at a sampling rate equivalent to a chiprate at which chips of a known signal are timed. The receiver signalsamples are segmented into receiver signal segments, which are filteredby respective matched filters to produce respective filter responsesequences. The matched filters are maximally responsive to respectiveknown signal segments segmented from the known signal. Indexes areassigned to elements of the filter response sequences to define an arraythereof. The elements of the filter response sequences are indexed in afirst row order defined by a first sequence order of the known signalsegments as distributed across the matched filters and in a first columnorder defined by a second sequence order in which the receiver signalsamples of the receiver signal segments are filtered. Frequencytransforms are computed of elements of the filter response sequencesindexed in respective columns of the array to produce respectivefrequency response sequences. Indexes are assigned to elements of thefrequency response sequences to define another array thereof. Theelements of the frequency response sequences are indexed in a secondcolumn order defined by a third sequence order in which the receiversignal samples of the receiver signal segments are filtered and in asecond row order defined by a fourth sequence order of the frequencyresponse sequences. Channel effects imparted on the radio signal arejointly estimated from characteristics of the other array at which atleast one local maximum is located.

BRIEF DESCRIPTION OF THE DRAWINGS

The descriptions herein are to be read in view of the following figures,where like reference numerals across figures refer to like functionalcomponents.

FIG. 1 is a diagram illustrating an example scenario in which principlesdescribed herein apply.

FIG. 2 is a schematic diagram of an example communication system and anassociated channel model for purposes of explaining certain fundamentalsand assumptions for an example receiver design embodying principles ofthis disclosure.

FIG. 3 is a schematic diagram of a matched filter bank scheme by whichprinciples of this disclosure can be embodied.

FIG. 4 is a schematic diagram of an estimator by which principles of thepresent disclosure can be embodied.

FIG. 5 is a schematic block diagram of example timing and gain trackingby which principles of this disclosure can be embodied.

FIG. 6 is a schematic block diagram of an example embodiment accordingto principles of this disclosure using a maximal-ratio combiner (MRC).

FIG. 7 is a schematic block diagram of a message bit detector by whichprinciples of the present disclosure may be embodied.

FIG. 8 is a schematic block diagram of an example receiver by whichprinciples of this disclosure can be embodied.

FIG. 9 is a flow diagram of an example receiver process by whichprinciples of the present disclosure can be embodied.

FIG. 10 is a schematic block diagram of example receiver infrastructureby which principles of the present disclosure can be embodied.

FIGS. 11A-11D depict a set of signals by which fundamentals of amodified technique embodying principles of the present disclosure isexplained.

FIG. 12 is a schematic block diagram of an estimator by which principlesof the present disclosure can be embodied.

FIG. 13 is a flow diagram of an example receiver process 1300 by whichprinciples of the present disclosure can be embodied.

DETAILED DESCRIPTION

The present inventive concept is best described through certainembodiments thereof, which are described in detail herein with referenceto the accompanying drawings, wherein like reference numerals refer tolike features throughout. The inventive concept is not limited to theillustrative embodiments described below and the following descriptionsshould be read in such light.

The word exemplary is used herein to mean, “serving as an example,instance or illustration.” Any embodiment of construction, process,design, technique, etc., designated herein as exemplary is notnecessarily to be construed as preferred or advantageous over other suchembodiments. Particular quality or fitness of the examples indicatedherein as exemplary is neither intended nor should be inferred.

Additionally, mathematical expressions are contained herein and thoseprinciples conveyed thereby are to be taken as being thoroughlydescribed therewith. It is to be understood that where mathematics areused, such is for succinct description of the underlying principlesbeing explained and, unless otherwise expressed, no other purpose isimplied or should be inferred. It will be clear from this disclosureoverall how the mathematics herein pertain to the inventive principlesand, where embodiment of the principles underlying the mathematicalexpressions is intended, the ordinarily skilled artisan will recognizenumerous techniques to carry out physical manifestations of theprinciples being mathematically expressed.

The figures described herein include schematic block diagramsillustrating various functional modules for purposes of description andexplanation. Such diagrams are not intended to serve as electricalschematics and interconnections illustrated are merely to depict signalflow, various interoperations between functional components and/orprocesses and are not necessarily direct electrical connections betweensuch components. Moreover, the functionality illustrated and describedvia separate components need not be distributed as shown, and thediscrete blocks in the diagrams are not necessarily intended to depictdiscrete electrical components.

FIG. 1 is a diagram illustrating an example scenario in which theprinciples described herein might apply. The broad goal in theillustrated example is communication between seagoing and airbornevessels 110 and 120, respectively. The illustrated scenario assumesstill waters and a low altitude flight path of the airborne vessel 120.Under such conditions, a transmitted information bearing signal, e.g.,from the seagoing vessel 110, becomes subject to multiple physicalprocesses or effects before it arrives at a receiver on the airbornevessel 120. For example, in addition to line-of-sight (LOS) propagation,copies of the transmitted signal that have reflected from the seasurface arrive at the receiver, indicated in FIG. 1 as multi-path (MP)propagation. In FIG. 1, signal peak 102 corresponds to LOS propagationof the transmitted signal and signal peaks 104 and 106 correspond to MPpropagation of the transmitted signal. The received signal 130 is acomposite of the LOS and MP signals. Physical processes assertthemselves on the LOS and MP signals, which can be modeled throughchannel gain coefficients h₁, h₂ and h₃, and arrival times τ₁, τ₂ and τ₃as illustrated in FIG. 1. Additionally, in the general case, there is asignificant difference in velocity between the two vessels and,accordingly, the transmitted signal undergoes a Doppler shift beforearriving at the receiver. Each component signal corresponding to thedifferent propagation paths may undergo a distinct Doppler shift e.g.,ω_(D,1), ω_(D,2) and ω_(D,3).

FIG. 2 is a schematic diagram of an example communication system 200with an associated channel model 250 by which certain fundamentals andassumptions for an example receiver design embodying principles of thisdisclosure is explained. Example communication system 200 comprises atransmitter 210 and a receiver 240, although only those components oftransmitter 210 and receiver 240 that are pertinent to channel model 250are illustrated in the figure. Communication system 200 is assumed to beoperating under the conditions described above with reference to FIG. 1,i.e., calm waters and low altitude flight path of the receivingaircraft. Transmitter 210 generates an information bearing signal thatis conveyed by electromagnetic wave propagation through a medium 215(such as air) to receiver 240.

In certain embodiments, communication system 200 implementsdirect-sequence spread spectrum (DSSS) modulation and, as such,transmitter 210 may include a spreading component 220 that takes as itsinput bits of a message signal and produces at its output radio pulsesof the frequency-spread signal that are timed at the chip rate (e.g.,135 MHz). As illustrated in FIG. 2, within spreading component 220, eachmessage bit a_(n) may be upsampled to a length of N and may besubsequently modulated with a pseudo-noise (PN) code s_(i) to producespread signal chips x_(i). In certain embodiments, each successivemessage bit a_(n) in the message sequence is modulated by a sequence ofN chips of the PN code s_(i) such that successive sequences of PN codechips modulate successive message bits a_(n). Chips x_(i) are providedto a digital-to-analog converter (DAC) 212 having a conversion rateequal to the chip rate. The resulting analog signal is upconverted tothe carrier frequency (e.g., 5 GHz) by upconverter 214 and the resultinginformation bearing signal is emitted into medium 215 through an antenna216.

On the side of receiver 240, the transmitted information bearing signal,which has been subjected to the physical phenomena described above, maybe intercepted by an antenna 246 and then downconverted by adownconverter 244. The downconverted signal may then be sampled at thechip rate by an analog-to-digital converter (ADC) 242 to produce complexsamples of the transmitted chips, denoted herein as receiver signalsamples r_(i).

Channel model 250 represents the primary physical processes by whicheach transmitted chip x_(i) is received as samples r_(i) and includesnot only atmospheric effects, but also system effects ofup/down-conversion, sampling, amplification (amplifiers notillustrated), etc. As illustrated in FIG. 2, each chip x_(i) is subjectto delays τ_(m)(i) that correspond to the length of the m-th propagationpath. In channel model 250, the transmitted signal propagating alongeach of the paths may undergo a Doppler shift ω_(D,m)(i), which isreflected in channel model 250 at 252 a-252 c for Doppler shiftsω_(D,1)(i), ω_(D,2)(i) and ω_(D,3)(i). Additionally, the propagationdelays associated with each propagation path are represented in channelmodel 250 at 254 a-254 c for respective delay times of τ₁(i), τ₂(i) andτ₃(i). Each chip x_(i) can also undergo one or more of amplification,range-dependent spreading, atmospheric attenuation, etc. that affect thereceived signal strength. These factors are reflected in channel model250 through channel gain coefficients h_(m)(i), which, in the exampleillustrated, are represented at 256 a-256 c for channel gaincoefficients h₁(i), h₂(i) and h₃(i). As discussed above, the receivedsignal is a composite of the multipath signal components and suchcomposition is reflected in channel model 250 as a summation indicatedat 258. The transmitted signal may also be subject to additive whiteGaussian noise (AWGN) and jamming signals, indicated at 259, which areameliorated by the spread spectrum technique.

To decode the transmitted message from the received signal, compensationfor the channel effects described above should be applied. To that end,one must know or at least estimate the number of propagation paths and,for each of those paths, the propagation delay τ_(m), the Doppler shiftω_(D,m) and the channel gain h_(m). The principles described in thisdisclosure include a technique to estimate these parameters withouthaving to resort to an expensive exhaustive search over all Dopplershift and delay possibilities.

FIG. 3 is a schematic diagram of a matched filter bank scheme by whichprinciples of this disclosure can be embodied. As discussed above, eachmessage bit a_(n) is modulated at transmitter 210 by N chips of the PNcode such that successive message bits are modulated by a correspondingone of successive N chips of the PN code. Accordingly, a filter bank 300of L matched filters 320 a-320 l, representatively referred to herein asmatched filter(s) 320, may be constructed such that each matched filter320 is matched for maximal response in terms of signal-to-noise ratio tothe particular sequence of N chips of the PN code that modulated eachcorresponding message bit a_(n). In FIG. 3, this action is illustratedas a path selector 305 that feeds successive matched filters 320 at themessage bit rate R_(b). That is, as is illustrated in the figure,matched filter 320 a receives a first sequence of N successive samplesr₀-r_(N−1) corresponding to a first message bit modulated by the first Nchips of the PN code, matched filter 320 b receives a second sequence ofN successive samples r_(N)-r_(2N−1) corresponding to a second messagebit modulated by the second N chips of the PN code, and so on, for eachof L matched filters 320. The output of each matched filter 320 is afilter response sequence λ_(i) ^((k)), 2N−1 samples of the correlationbetween the k-th received message bit (delayed, Doppler shifted and withchannel gain applied) and the corresponding k-th successive sequence ofN PN code chips used to modulate that message bit.

FIG. 4 is a schematic diagram of an estimator 400 by which principles ofthe present disclosure can be embodied. Estimator 400 may take as itsinput a sequence of complex receiver signal samples r_(i) from areceiver front end 405, such as by the technique described above withreference to FIG. 2. Samples r_(i) may be provided to a firstserial-to-parallel (S/P) converter 410 that assembles the sequence r_(i)into L successive sequences of N successive samples r_(i) (vectorsr_(n)) and may provide those data sets to respective L matched filtersof matched filter bank 415. Each of the matched filters in filter bank415 may be maximally responsive to a corresponding one of successivesequences of N successive chips of the PN code, i.e., the matched filterfiltering a first sequence of receiver signal samples r_(i) may bemaximally responsive to a first N chips of the PN code sequence, asecond sequence of receiver signal samples r_(i) may be maximallyresponsive to a second N chips of the PN code sequence, and so forth.The outputs of matched filter bank 415 may be L 2N−1 point filterresponse data sequences (vectors λ_(n)), which may be provided to asecond S/P converter 420. S/P converter 420 may be constructed orotherwise configured to align the L 2N−1 point filter response datasequences such that the sequence elements thereof at like sequencelocations (relative to the first data point of each filter response datasequence) across the filter response data sequences collectively formrespective temporal data sequences timed at the message bit rate. Thismay be achieved by constructing an L×(2N−1) array or matrix datastructure in memory, referred to herein as the Λ-matrix 425, having Lrows corresponding to the L matched filters and 2N−1 columnscorresponding to the 2N−1 point filter response data sequences. Eachcolumn in the Λ-matrix 425 may be comprised of elements of the filterresponse sequences at like sequence locations thereof that arethemselves sequential in terms of the succession order in the PN code ofthe NPN code chips with which sequential message bits are modulated andto which the respective matched filters are maximally responsive.

In accordance with principles of this disclosure, each column ofΛ-matrix 425 or, alternatively, elements of the filter responsesequences at like sequence locations thereof across the filter responsesequences, may be provided to one of 2N−1 fast Fourier transform (FFT)processors, representatively illustrated by FFT processor 430. Each FFTprocessor 430 computes the FFT for each column of Λ-matrix 425 based onthe message bit rate Rb. The resulting frequency spectrum from each FFTprocessor 430 may be stored in memory as a column in an L×(2N−1) arrayor matrix data structure 435, referred to herein as the Ψ-matrix 435.When properly constructed, Ψ-matrix 435 represents a parallel searchover a parameter space comprising L different possible frequency offsets(rows of Ψ-matrix 435) and 2N−1 different delays (columns of Ψ-matrix435). The correct frequency offset and delay may manifest as an entry inΨ-matrix 435 having a local maximum. Such peaks are illustrated in theΨ-matrix example 450.

A peak detector 440 may be employed to seek peaks (local maxima) in thedata of Ψ-matrix 435, where multiple peaks represent multiplepropagation paths. The coordinates of each peak in Ψ-matrix 435 is ajoint estimate of the path-specific delay and the path-specific Dopplershift for the corresponding propagation path while the magnitude of eachpeak represents, within a constant value and noise contributions, thechannel gain for the corresponding propagation path. The results of thisanalysis reveal the number of propagation paths (from the number ofpeaks), estimates of the path delays {circumflex over (τ)}_(m) (from theΨ-matrix column coordinate of the corresponding peak), estimates of theDoppler shifts {circumflex over (ω)}_(D,m) (from the Ψ-matrix rowcoordinate of the corresponding peak) and estimates of the channel gainĥ_(m). According to one or more embodiments, channel gain estimates mustbe obtained from the Ψ-matrix before the magnitude of the matrix istaken, since h_(m) is normally a complex number. The channel gainestimates may be formulated from

${\hat{h}}_{m} = \frac{\Psi_{m}}{e^{j\; {\hat{\omega}}_{{D,m}\;}{\hat{\tau}}_{m}}{\sum_{l = 0}^{L - 1}{e^{{- j}\; {\hat{\omega}}_{D,m}{lN}}{A^{({l,l})}\left\lbrack {0,{\hat{\omega}}_{D,m}} \right)}}}}$

where Ψ_(m) corresponds to the entry of the Ψ-matrix for thecorresponding peak before the magnitude is taken, andA^((l,l))[i,ω_(D))=x_(i) ^((l))e^(jω) ^(D) ^(i)*x_(−i) ^(*(l)) wherex_(i) ^((l)) are the chips that correspond to the lth bit of transmittedsignal. The estimates {circumflex over (τ)}_(m) and {circumflex over(ω)}_(D,m) may be used for coarse signal acquisition, as explainedbelow, while estimates ĥ_(m) may be used to initialize the adaptivechannel estimators, as described below.

The number L of matched filters in filter bank 415 may be a selectableparameter that may correspond to the number of message bits used forcoarse acquisition of the received signal. If L is chosen too small, thecoarse acquisition process is susceptible to noise and, since thefrequency step of the FFTs is R_(b)/L, frequency resolution will belost. If L is chosen too large, agility for rapidly changing channelswould be lost.

The coarse channel effects estimation illustrated in FIG. 4 may beperformed everytime the receiver loses synchronization with thetransmitter. For example, it may be performed at the start of every“fresh” transmission (in which the receiver has no present knowledge ofthe channel state) or whenever the receiver loses track of the channelstate.

FIG. 5 is a schematic block diagram of example timing and gain trackingby which principles of this disclosure can be embodied. The timing andgain tracking utilizes the estimates of {circumflex over (τ)}_(m),{circumflex over (ω)}_(D,m) and ĥ_(m) for coarse signal acquisition andrefines such acquisition through suitable receiver mechanisms, examplesof which are described herein.

As illustrated in FIG. 5, complex samples r_(i) may first be provided toa carrier phase shifting component 505 that compensates for carrierphase differences through a variable phase shift θ_(i). Example detailsfor example carrier phase tracking are provided below. The output ofcarrier phase shifting component 505 may be provided to a Doppler/delayprocessing component 550, which will be referred to herein as DDPcomponent 550. As will be further demonstrated below, a DDP component550 may be instantiated for each propagation path discovered by theestimation technique described above with reference to FIG. 4.

The signal r_(i)e^(−l θ) ^(i) may first be provided to a Dopplercompensation component 510 within DDP component 550. Dopplercompensation component 510 may apply the path dependent Doppler shiftestimate {circumflex over (ω)}_(D,m) to generate a phase compensatedsignal w_(i). Phase compensated signal w_(i) may then be provided to aset of programmable or otherwise variable delay lines 552 a-552 c,representatively referred to herein as delay line(s) 552. Delay lines552 form part of a delay locked loop and, to that end, one of delaylines 552, delay line 552 b for example, is fixedly advanced by half achip period while another of delay lines 552, delay line 552 c forexample, is fixedly retarded by half a chip period. As illustrated inthe figure, delay lines 552 generate three (3) signals: v_(i), which areprogrammatically delayed by an amount equal to the path-dependent delayestimate {circumflex over (τ)}_(m) plus a variable refining amountd_(i),

${v\begin{matrix}{EARLY} \\i\end{matrix}},$

which is programmatically delayed by an amount equal to {circumflex over(τ)}_(m)+d_(i) plus the advance amount of 0.5 times the chip period, and

${v\begin{matrix}{LATE} \\i\end{matrix}},$

which is programmatically delayed by an amount equal to {circumflex over(τ)}_(m)+d_(i) minus the hold back amount of 0.5 times the chip period.The three signals v_(i),

$v\begin{matrix}{EARLY} \\i\end{matrix}\mspace{14mu} {and}\mspace{14mu} v\begin{matrix}{LATE} \\i\end{matrix}$

may be provided to respective despreading components 554 a-554 c,representatively referred to herein as despreading component(s) 554,where the received signal is despread using the PN code with which thetransmitted signal was frequency-spread. The outputs of despreadingcomponents 554 are timed at the message bit rate Rb and are denotedherein as y_(n),

$y\begin{matrix}{EARLY} \\n\end{matrix}\mspace{14mu} {and}\mspace{14mu} y\begin{matrix}{LATE} \\n\end{matrix}$

in correspondence with the input signals v_(i),

$v\begin{matrix}{EARLY} \\i\end{matrix}\mspace{14mu} {and}\mspace{14mu} v\begin{matrix}{LATE} \\i\end{matrix}$

from which they are derived. The signal y_(n) is forwarded to channelgain compensation, which will be discussed below. However, the signals

$y\begin{matrix}{EARLY} \\n\end{matrix}\mspace{14mu} {and}\mspace{14mu} y\begin{matrix}{LATE} \\n\end{matrix}$

serve as input signals to the delay locked loop and, as such, aredelivered to a timing error detector (TED) 556. The timing error signalgenerated by TED 556 may follow the relation

$ɛ_{n} = {{{y\begin{matrix}{EARLY} \\n\end{matrix}}} - {{y\begin{matrix}{LATE} \\n\end{matrix}}}}$

and may be used to advance or delay a phase-locked loop (PLL) 558. PLL558 generates a signal that is indicative of a variable delay amountd_(m)[n] that is commensurate with the timing error ε_(n) for n-thmessage bit period. The delay amount d_(m)[n] may be adjusted to a fineresolution, including subchip intervals, the timing tracking resolutionbeing limited only by the PLL used. The PLL output signal that isindicative of d_(m)[n] may be applied across the set of delay lines 552to close the delay locked loop. In one embodiment, the delay locked loopis updated at the bit rate R_(b) according to:

d_(m)[n + 1] = d_(m)[n] + α ɛ_(n) + βΣ_(i = 0)^(n)ɛ_(i),

where α and β are predetermined weights. It is to be understood that theprinciples described herein are not limited to this second orderexample.

The signal y_(n) may be generated at the message bit rate R_(b) by DDPcomponent 550 from carrier phase adjusted received signal samplesr_(i)e^(−jθ) ^(i) provided to DDP component 550 at the chip rate. Asstated above, DDP output signal y_(n) may be provided to channel gainestimation and tracking (CGET) component 560 by which the channel gainestimates ĥ_(m) are compensated. As illustrated in FIG. 5, CGETcomponent 560 may comprise a variable gain component 520 and anestimator component 530. In one embodiment, estimator component 530computes channel gain estimates according to

ĥ _(m)(n+1)=(1−y)ĥ _(m)(n)+γy _(n)a*_(n),

where γ is a training weight and a_(n) ^(*) is the complex conjugate ofthe n-th message bit as recovered at the receiver. The channel gainestimates ĥ_(m) may be used to initialize this second order estimationprocess. The complex conjugate of the channel gain updates ĥ_(m)(n+1)are provided to variable gain component 520 to produce thepath-dependent compensated signal ĥ_(m) ^(*)y_(n). For decision directedoutput, a_(n) can be replaced by â_(n)=sign(Re{ĥ_(m) ^(*)y_(n)}).

FIG. 6 is a schematic block diagram of an example embodiment using amaximal-ratio combiner (MRC) 610 that combines the path-specific signalsdescribed above to yield a signal z_(n) for which the signal-to-noiseratio is maximized. As illustrated in the figure, the signal ĥ*_(m)y_(n)may be produced by each of a set of finger components 620 a-620 c,representatively referred to herein as finger component(s) 620, and maybe provided to a summing component 630. Each finger component 620, asrepresentatively depicted in finger component 620 a, may comprise a DDPcomponent 550 and a CGET component 560, each of which may operate in amanner described above. Summing component 630 accepts the Doppler, delayand channel gain compensated signals and combines them. MRC combiner 610controls each CGET 560 so that the gain of each channel is proportionalto the root mean square (RMS) signal level and inversely proportional tothe RMS noise level in that channel.

FIG. 7 is a schematic block diagram of a message bit detector 700 bywhich principles of the present disclosure may be embodied. Message bitdetector 700 includes finger components 620 and summing component 630 asdescribed above, as well as a carrier phase tracking component 740 and adecision component 710. Decision component 710 may make a decision as tothe state of the received message bit estimate â_(n). In the exampleembodiment, where the message bits a_(n)∈{−1,1}, decision component 710may be realized by evaluating the arithmetic sign of the rake combinedsignal z_(n), e.g., â_(n)=sign(Re{z_(n)}).

As illustrated in FIG. 7, carrier phase tracking component 740 maycomprise a phase detector component 720 and a PLL 730. Phase detectorcomponent 720 may determine a phase error, such as by

${\hat{\varphi}}_{n} = {{\sin^{- 1}\left( {{Im}\left\{ \frac{y_{n}a_{n}^{*}}{y_{n}} \right\}} \right)}.}$

Based on the error signal {circumflex over (ϕ)} _(n), PLL 730 maygenerate a signal indicative of a compensation angle according to

${{\overset{\hat{}}{\theta}\left( {n + 1} \right)} = {{\overset{\hat{}}{\theta}(n)} + {\rho {\overset{\hat{}}{\varphi}}_{n}} + {{\xi\Sigma}_{i = 0}^{n}{\overset{\hat{}}{\varphi}}_{i}}}},$

where ρ and ξ are predetermined weights. The PLL output signal may beprovided to carrier phase compensation component 505, which applies thephase compensation angle {circumflex over (θ)}_(i) to the receivedsamples r_(i) to produce the signal r_(i)e^(−j{circumflex over (θ)})^(i) provided to each of the finger components 620.

FIG. 8 is a schematic block diagram of an example receiver 800 by whichprinciples of this disclosure can be embodied. Receiver 800 may comprisean estimator component 820 and a detector component 850 that operate ina manner similar to analogous components described above. However,example receiver 800 has been modified for applications in which theDoppler shift ω_(D) differs very little from propagation path topropagation path. In such a case, the channel can be modeled with asingle Doppler shift ω_(D) applied on the composite signal as opposed tobeing applied on a per-path basis. This is reflected in the channelmodel 810. Accordingly, as illustrated in the figure, a single Dopplerestimate {circumflex over (ω)}_(D) may be generated by estimatorcomponent 820 and the Doppler compensation is removed from each fingercomponent 830 a-830 c and replaced by a single Doppler compensationcomponent 840 that precedes the rake combining. Outside thismodification, the operations performed by receiver 800 are substantiallythe same as those previously described.

FIG. 9 is a flow diagram of an example receiver process 900 by whichprinciples of the present disclosure can be embodied. In operations 902and 904, tracking indexes i, j are initialized to one (1). In operation906, complex receiver signal samples are received at the chip rate. Inoperation 908, a receiver signal sample may be added to a μ_(n) vector,which is a temporary vector of receiver signal samples. In operation910, the index j is incremented and, in operation 912, it is compared toN, the number of chips per message bit. If index j is less than N,process 900 returns to operation 908 and continues from that point.

If index j is equal to N, process 900 may transition to operation 914,whereby the N complex receiver samples in vector μ_(n) may be filteredby a matched filter that is maximally responsive to the j-th sequence ofN chips of the PN code used to spread the transmitted signal. Inoperation 916, the 2N−1 filtered samples are added as the j-th vectorμ_(n) of the Λ-matrix 425. In operation 918, the index i may beincremented and in operation 920, index i may be compared to L, thenumber of message bits used for coarse signal acquisition. If index i isless than L, process 900 may return to operation 904 and continues fromthat point.

If index i is equal to L, operation 900 transitions to operation 922, bywhich message bit rate based FFTs are executed on respective columns ofthe Λ-matrix 425 to produce corresponding columns of the Ψ-matrix 435.The Ψ-matrix 435 contains elements of the frequency response sequences(generated by the FFTs) that are indexed in a column order defined bythe sequence location of the elements of the filter response sequenceson which the respective frequency transforms were computed and in a roworder defined by a succession order of the sequences of chips of the PNcode to which the respective matched filters that produced therespective filter response sequences (on which the FFTs are performed)are maximally responsive.

In operation 924, coordinates of the peaks (local maxima) in theΨ-matrix 435 are determined. In operation 926, the number of propagationpaths is determined from the number of peaks in the Ψ-matrix 435. Inoperation 928, a tracking index k is initialized to one (1). Inoperation 930, a Doppler shift estimate of the k-th propagation path isdetermined from the row coordinate of the k-th peak. In operation 932, apropagation delay estimate corresponding to the k-th propagation path isdetermined from the column coordinate of the k-th peak. In operation934, a channel gain estimate is determined from the magnitude of thek-th peak. In operation 936, it is determined whether k is equal to thenumber of peaks M in the Ψ-matrix and, if not, process 900 maytransition to operation 938, by which the k index is incremented.Process 900 may then transition to operation 930 and continues from thatpoint.

If index k is equal to the number of peaks in the Ψ-matrix, then process900 may transition to operation 940, by which a number of fingers of arake combiner equivalent to the number of peaks is instantiated in areceiver detector. In operation 942, the Doppler and delay estimates areprovided to the receiver detector for coarse signal acquisition. Inoperation 944, the adaptive channel gain estimator (embodied by CGETs,for example) is initialized with the channel gain estimates. Inoperation 946, message bits are generated by the receiver detectorthrough coarse signal acquisition using the Doppler shift estimates, thepropagation delay estimates and the channel gain estimates.

FIG. 10 is a schematic block diagram of example receiver infrastructure1000 by which principles of the present disclosure can be embodied. InFIG. 10, receiver infrastructure 1000 is abstracted through specificfunctional modules, each of which leverages circuit technologies andarchitectures that may be specific to the functions being performed bythat module. It will be appreciated by those skilled in the art that thedivision of the receiver 1000 into functional modules or blocks asillustrated in FIG. 10 is conceptual and is not intended to indicatehard boundaries of functionality or physical groupings of components. Inpractice, the illustrated functional modules may be combined, divided,and otherwise repartitioned into other modules without deviating fromthe scope and spirit of the present inventive concept.

Radio circuitry 1010 may comprise circuitry constructed or otherwiseconfigured for processing high-frequency signals. Among other things,radio circuitry 1010 may implement a radio front-end that interceptsradio-frequency signals (e.g., 5 GHz) and generates therefrom a lowerfrequency receiver signal for information extraction. To that end, radiocircuitry 1010 may include one or more antennas, low-noise amplifiers,filters, mixers, and other components (e.g., an isolator ortransmit-receive switch) coupled one with the others (e.g., bytransmission lines, electromagnetic couplers, general conductor pathsand the like) to define suitable radio-frequency signal processingpaths. The principles described in this disclosure are not limited toparticular radio configurations so long as the information conveyed inthe signal intercepted and processed thereby is recoverable using theforegoing techniques or their equivalents.

Analog-to-digital (A2D) circuitry 1040 may comprise circuitryconstructed or otherwise configured for sampling the receiver signalproduced by radio circuitry 1010. In certain embodiments, the samplingrate at which receiver signal samples are produced is equivalent to thechip rate (e.g., 135 MHz). A2D circuitry 1040 may generate complexnumbers in a known machine-readable format corresponding to the state ofthe receiver signal at that sampling period. Numerous different A2Darchitectures may be used in conjunction with the principles of thisdisclosure, as the skilled artisan will appreciate and acknowledge.

Memory circuitry 1050 may comprise circuitry constructed or otherwiseconfigured for storing various data items, including the complexreceiver signal samples produced by A2D circuitry 1040. In addition tostoring individual digital data items, memory circuitry 1050 may includememory management circuitry and components by which various datastructures, e.g., vectors and matrixes, are efficiently stored andretrieved for vector and matrix processing. Additionally, memorycircuitry 1050 may include circuitry constructed or otherwise configuredfor storing and retrieving processor instruction code for one or moreprocessors, where the processor instruction code may implement, throughexecution by a suitable processor, one or more of the features of theestimator and detector described above, e.g., matched filters, FFTprocessors, peak detector, rake finger processors, MRC, etc. Memorycircuitry 1050 may store system variables and data processing parametersas well. Memory circuitry 1050 may include both persistent memorycomponents, e.g., flash drive, hard disk drive, read-only memory, etc.as well as volatile memory, such as random access memory. Multiplememory technologies and techniques may be used in conjunction with theprinciples described herein without departing from the spirit andintended scope thereof.

Digital signal processing (DSP) circuitry 1020 may include circuitryconstructed or otherwise configured for processing complex digitalsignals, such as the receiver signal samples produced by A2D circuitry1040 and stored in memory circuitry 1050. Such processing may proceedaccording to DSP instruction code stored in memory circuitry 1050 andmay include operations such as addition, subtraction, multiplication anddivision of complex numbers, vector and matrix arithmetic, complexmodulation, frequency transforms (e.g., FFTs), matched filtering, peakdetection, data delays, signal despreading, timing error detection, PLLemulation, message bits decoding, etc., along with various otheroperations and algorithms discussed herein. DSP circuitry may includecommercially-available programmable digital signal processors and/orother circuitry constructed or otherwise configured to perform variousdigital signal processing tasks. Such other circuitry may be constructedfrom programmable logic components (e.g., field programmable gatearrays), application specific integrated circuits, discrete logic andother circuit components. In certain embodiments, the operatingfrequency of DSP circuitry 1020 exceeds the sampling frequency at whichreceiver signal samples are generated.

General processing, control and interface (GPCI) circuitry 1030 mayinclude circuitry constructed or otherwise configured for executingcontrol, coordination and interface tasks. In one embodiment, GPCIcircuitry 1030 implements a user interface through which a user canreceive messages (decoded message bits) and modify various systemcontrol parameters. GPCI circuitry 1030 may also implement anapplication programming interface (API) through which external devicesmay remotely operate receiver infrastructure 1000. GPCI circuitry 1030may include programmable logic, (e.g., microprocessors,microcontrollers) and human-machine interface devices (HMIDs) includingdisplays, keyboards, mice, earphones, microphones, etc.

Clock circuitry 1060 may include circuitry constructed or otherwiseconfigured for generating timing signals for receiver infrastructure1000. In one embodiment, clock circuitry 1060 comprises a masteroscillator from which all other timing signals (e.g., message bit rateclock, chip rate clock) are derived. Clock circuitry 1060 may include astable local oscillator circuit and/or a coherent oscillator fordown-converting the received information bearing signal into thereceiver signal that is sampled. Clock circuitry 1060 may be constructedor otherwise configured to provide processor operating clock signals toDSP circuitry 1020 and GPCI circuitry 1030.

Receiver infrastructure 1000 may include bus circuitry 1070 constructedor otherwise configured for conveying the various signals describedherein from one component to another. As such, bus circuitry 1070 maycomprise conductive structures forming data and control signal pathsthat may have multi-conductor configurations corresponding with the dataor control word length.

The techniques described above may be extended or otherwise modified toincrease the range over which the estimated propagation delay iscomputed. To achieve this range expansion, a maximum expected delayτ_(max) may be specified such that computed delay estimates fall withinthe range {−τ_(max), . . . , 0, . . . τ_(max)}.

FIGS. 11A-11D, collectively referred to herein as FIG. 11, depict a setof signals by which fundamentals of the modified technique can beexplained. FIG. 11A illustrates an arbitrary signal s[i], written ass_(i) herein, that is known to the receiver. As illustrated in thefigure, signal s_(i) is L chips long and, in certain embodiments, istransmitted from a transmitter for receiver training or specifically forchannel estimation.

In certain embodiments, signal s_(i) is partitioned or otherwisesegmented into K segments {s _(i) ⁽⁰⁾

_(i) ^((k)), . . . ,s_(i) ^((K−1))}, each including P chips of s_(i)such that

$P = {\frac{K}{L}.}$

Such segmentation may be achieved through suitable windowing, such as bythe window w_(i) illustrated in FIG. 11B, so that s_(i)^((k))=S_(kP+i)w_(i). When so segmented,

$s_{i} = {\sum_{k = 0}^{K - 1}{s_{i - {k\frac{L}{K}}}^{(k)}.}}$

In accordance with the principles of the present disclosure, a filterbank of K matched filters may be constructed or otherwise configuredsuch that each of the K matched filters is maximally responsive to acorresponding one of the segments s_(i) ^((k)). With similarity to thetechnique described above, the output of each matched filter may beassigned to a corresponding row of a matrix and a frequency transform,e.g., an FFT, may be performed on each column of that matrix. Furtheraspects of these operations are described below.

It is to be noted that there is a performance tradeoff in the choice ofP and, as such, P can be optimized based on the application of thepresent inventive concepts. A smaller P, and consequently a larger K,results in more FFT bins which allows the receiver to estimate largerDoppler frequencies without aliasing. Additionally, a smaller P reducesthe amount of Doppler loss at the output of each filter resulting inmore prominent peaks in the output waveform. On the other hand, a smallP results in high computational complexity (measured by the number ofmultiply/add operations the receiver must execute) as K would be largerand the column-wise FFT would be K point.

FIG. 11C depicts a signal r_(i), which, as before, is the receivedversion of s_(i) transmitted by the transmitter, conveyed through thechannel medium, and sampled at the chip rate by the receiver. Excluding,for the moment, consideration of channel gain and Doppler shift solelyto simplify the explanation of the present embodiment in terms ofpropagation delay, the received signal r_(i) can be expressed asr_(i)=s_(i−τ)+η_(i), where τ is the propagation delay τ∈T={−τ_(max)

_(max)} being sought and η_(i) is noise. One search strategy correlatesthe received signal with s_(i−d) for all possible candidate delays d andthen selects the value of d that resulted in the highest correlation.For the case of additive white Gaussian noise (AWGN), this strategyproduces the maximum-likelihood estimate of τ. Benefits of efficiencycan be had by segmenting the received signal into K segments r_(i)^((k)), passing each r_(i) ^((k)) through a filter matched to s_(i)^((k)) and indicating which filter of the K filters exhibited thegreatest response. As in the example described above, this strategy maybe realized by suitable circuitry, including programmable circuitry,capable of generating an output signal λ_(i) for a given input signalr_(i) by way of a filtering operation λ_(i)=Σ_(k=0) ^(K−0)r_(i)^((k))*s_(−i) ^((k)), where s_(i) is the known signal and “*” is theconvolution operator. This filtering operation can be implemented by afilter bank of K parallel filters respectively matched to segments ofthe known signal s_(i).

If λ_(i) were to be computed for all i, the input to the k-th matchedfilter would have to be

${r_{i}^{(k)} = r_{{k\frac{L}{K}} + i}},$

a delayed/advanced version of the entire received signal. However,according to principles described herein, λ_(i) may only be evaluatedfor values of i that correspond to propagation delays within T={−τ_(max)

_(max)}. Thus, the input r_(i) ^((k)) provided to the k-th matchedfilter can be limited or otherwise bound to contain only the relevantsamples. This may be achieved by a windowing operation such that

${r_{i}^{(k)} = {r_{{k\frac{L}{K}} + i}w_{i}^{\prime}}},$

where w′_(i) is the window function. In certain embodiments, w′_(i) maybe a rectangular window for which w′_(i)=1 for values of i thatcorrespond to propagation delays within

$\mspace{20mu} {T^{\prime} = \left\{ {{{- \tau_{\max}}\text{?}\frac{L}{\text{?}}} + \tau_{\max} - 1} \right\}}$?indicates text missing or illegible when filed

and w′_(i) =0 elsewhere, as illustrated in FIG. 11D. Thus, the receivedsignal r_(i) is segmented into K segments, each of which being providedto a corresponding one of the K matched filters. It is to be observed inFIGS. 11A and 11C, that while s_(i) and r_(i) are both decomposed into Ksignal segments, the signal segments s_(i) ^((k)) are disjoint while thesegments r_(i) ^((k)) overlap.

By the technique described in the foregoing paragraphs, the matchedfilter bank output λ_(i) for values of i corresponding to τ└T can becomputed by segmenting the received sequence into K overlapping segmentsr_(i) ^((k)) through r_(i) ^((K−1)), passing the k-th segment through afilter matched to s_(i) ^((k)) and adding the results. At the output ofthe k-th matched filter,

$\lambda_{i}^{(k)} = {r_{{k\frac{L}{K}} + i}*{s_{- i}^{(k)}.}}$

For purposes of additional analysis by which the benefits of principlesdescribed herein can be further explained, the expression for λ_(i)^((k)) for values of i corresponding to τ∈T can be rewritten, after somemanipulation, as

${\lambda_{i}^{(k)} = {{R^{({k,k})}\left( {i - \tau} \right)} + {\sum_{j \neq k}{R^{({k,j})}\left( {i - \tau - {\left( {k - j} \right)\frac{L}{K}}} \right)}} + \eta_{i}^{(k)}}},$

where η_(i) ^((k)) is filtered noise and R^((k,j))(x)=Σv=−∞^(∞)s_(v+x)^(k)s_(v) ^(j) is the crosscorrelation function between the k-th signalsegment s_(i) ^((k)) and the x-th signal segment s_(i) ^((x)). Underthis definition, R^((k,k))(x) is the autocorrelation of the k-th signalsegment. While the summation limits on the crosscorrelation function areinfinite to emphasize that it depends on the entire signal segment,there will in fact be only L/K non-zero terms in the sum, i.e., for i∈{

/K−1}.

In the expression for λ_(i) ^((k)) given above, many of the terms in thesum Σ_(j≠k)(⋅) will be zero. For example, because each signal segments_(i) ^((k)) is L/K chips long, the crosscorrelation R^((k,j)) (x) willbe zero whenever the lag exceeds this length, i.e., |x|≥L/K. In the casewhen τ_(max)≤L/K, the expression for λ_(i) ^((k)) simplifies to

$\lambda_{i}^{(k)} = {{R^{({k,k})}\left( {i - \tau} \right)} + {R^{({k,{k - 1}})}\left( {i - \tau + \frac{L}{K}} \right)} + {R^{({k,{k + 1}})}\left( {i - \tau - \frac{L}{K}} \right)} + {\eta_{i}^{(k)}.}}$

Here, the first term will often dominate, especially when i is near τ,where R^((k,k−1)) and R^((k,k+1)) compute the correlation between twopotentially unrelated signal segments that barely overlap and R^((k,k))computes the correlation of a signal with a barely delayed version ofitself. In the extreme case when i=τ, the second and third terms areidentically zero.

When constructed or otherwise configured according to principles setforth in the preceding paragraphs, the impulse response of each matchedfilter in the filter bank is of length L/K and each input signal segmentis of length L/K+2τ_(max). Direct convolution would lead to an outputsignal of each matched filter that is of length 2L/K+2τ_(max)−1.However, embodiments of principles described herein require onlyD=2τ_(max)+1 chips to estimate τ, i.e., those whose index i correspondswith τ∈T={−τ_(max)

_(max)}. When so embodied on an infrastructure similar to thatillustrated in FIG. 8, the Λ-matrix 425 is a K×D matrix whose k-th row(for k∈{0,1, . . . ,K−1}) is λ_(i) ^((k)) for values of i thatcorrespond with τ∈T.

FIG. 12 is a schematic block diagram of an estimator 1200 by which thepresent disclosure can be embodied. The operational flow through theexample functional units illustrated in FIG. 12, with a few notablemodifications, is the same as that described with reference to FIG. 4.Indeed, estimator 400 of FIG. 4 and estimator 1200 of FIG. 12 are bothexample implementations of broader principles conveyed by thisdisclosure. In view of their similarities, the description below ofestimator 1200 will largely omit details of functionality that has beenpreviously described and that are implemented in estimator 1200 in likemanner as that implemented in estimator 400.

As illustrated in FIG. 12, complex receiver signal samples r_(i) from areceiver front end 405 may be provided to a segmenting component 1210that assembles the sequence r_(i) into K successive sequences ofP+2τ_(max) successive samples r_(i) (vectors r_(k)) based on a value ofτ_(max) provided thereto by a system administrator or by an externalsystem configured for estimating a maximum expected propagation delay.Each of data sets r_(k)may be provided to respective K matched filtersof matched filter bank 1215. Each of the matched filters in filter bank1215 may be maximally responsive to a corresponding one of successivesequences of P successive chips of the known signal s_(i), i.e., thematched filter filtering a first sequence of receiver signal samplesr_(i) may be maximally responsive to a first P chips of s_(i), thematched filter filtering a second sequence of receiver signal samplesr_(i) may be maximally responsive to a second P chips of s_(i), and soforth. The outputs of matched filter bank 1215 may be K 2P+2τ_(max)−1point filter response data sequences (vectors λ_(k)), which may beprovided to a second segmenting component 1220. Segmenting component1220 may be constructed or otherwise configured to limit vectors λ_(k)to the D=2τ_(max)+1 chips corresponding with τ∈T={−τ_(max)

_(max)} and to align the D point filter response data sequences in theK×D A-matrix, having K rows corresponding to the K matched filters and Dcolumns corresponding to the D point filter response data sequences.Each column in the Λmatrix may be comprised of elements of the filterresponse sequences at like sequence locations thereof that arethemselves sequential in terms of the succession order in s_(i).

In accordance with the principles of this disclosure, each column ofΛ-matrix 1225 or, alternatively, elements of the filter responsesequences at like sequence locations thereof across the filter responsesequences, may be provided to one of D fast Fourier transform (FFT)processors, representatively illustrated by FFT processor 1230. Each FFTprocessor 1230 computes the FFT for each column of Λ-matrix 1225 and theresulting frequency spectrum from each FFT processor 1230 may be storedin memory as a column in the K×D T-matrix 1235. As in the example ofFIG. 4, Ψ-matrix 1235 represents a parallel search over a parameterspace comprising K different possible frequency offsets (rows ofΨ-matrix 1235) and D different delays (columns of Ψ-matrix 1235). Thecorrect frequency offset and delay may manifest as an entry in Ψ-matrix1235 having a local maximum, which may be identified by peak detector440. As before, the number of propagation paths (from the number ofpeaks), estimates of the path delays {circumflex over (τ)}_(m) (from theΨ-matrix column coordinate of the corresponding peak), estimates of theDoppler shifts {circumflex over (ω)}_(D,m) (from the Ψ-matrix rowcoordinate of the corresponding peak) and estimates of the channel gainĥ_(m) (from

${\hat{h}}_{m} = \frac{\Psi_{m}}{e^{j{\hat{\omega}}_{D,m}\tau_{m}}{\sum_{k = 0}^{K - 1}{e^{{- j}{\hat{\omega}}_{D,m}{kP}}{A^{({k,k})}\left\lbrack {0,{\hat{\omega}}_{D,m}} \right)}}}}$

where A^((k,k))[i,ω_(D))=x_(i) ^((k))e^(jω) ^(D) ^(i)*x_(−i) ^(*(k)),and x_(i) ^((k)) is the kth segment of the transmitted signal) may bedetermined and provided to a receiver signal detector, such as detectorcomponent 850 illustrated in FIG. 8, where r_(i) at the signal detectoris that of an information bearing signal as opposed to the known signalprovided to estimator 1200.

FIG. 13 is a flow diagram of an example receiver process 1300 by whichprinciples of the present disclosure can be embodied. In operations 1302and 1304, tracking indexes i, j are initialized to one (1). In operation1306, complex receiver signal samples are received at the chip rate. Inoperation 1308, a receiver signal sample may be added to a μ_(n) vector,which is a temporary vector of receiver signal samples. In operation1310, the index j is incremented and, in operation 1312, it is comparedto P+2τ_(max), the number of chips that are to be provided to thematched filter bank. If index j is less than P+2τ_(max), process 1300returns to operation 1308 and continues from that point.

If index j is equal to P+2τ_(max), process 1300 may transition tooperation 1314, whereby the P+2τ_(max) complex receiver samples invector μ_(k) may be filtered by a matched filter that is maximallyresponsive to the j-th sequence of P chips of the known transmittedsignal. In operation 1316, λ_(j), the output of the j-th matched filter,is length-limited to D chips corresponding to {−τ_(max), . . . ,τ_(max)}. In operation 1318, the D samples of the j-th vector λ_(j) arestored as the j-th row of the Λ-matrix. In operation 1320, the index imay be incremented and in operation 1322, index i may be compared to K,the number of matched filters in the filter bank. If index i is lessthan K, process 1300 may return to operation 1304 and continue from thatpoint.

If index i is equal to K, operation 1300 transitions to operation 1324,by which FFTs are executed on respective columns of the Λ-matrix toproduce corresponding columns of the Ψ-matrix. The Ψ-matrix containselements of the frequency response sequences (generated by the FFTs)that are indexed in a column order defined by the sequence location ofthe elements of the filter response sequences on which the respectivefrequency transforms were computed and in a row order defined by asuccession order of the sequences of chips of the known transmittedsignal to which the respective matched filters that produced therespective filter response sequences (on which the FFTs are performed)are maximally responsive.

In operation 1326, coordinates of the peaks (local maxima) in theΨ-matrix are determined. In operation 1328, the number of propagationpaths is determined from the number of peaks in the Ψ-matrix. Inoperation 1330, a tracking index k is initialized to one (1). Inoperation 1332, a Doppler shift estimate of the k-th propagation path isdetermined from the row coordinate of the k-th peak. In operation 1334,a propagation delay estimate corresponding to the k-th propagation pathis determined from the column coordinate of the k-th peak. In operation1336, a channel gain estimate is determined from the magnitude of thek-th peak. In operation 1338, it is determined whether k is equal to thenumber of peaks M in the Ψ-matrix and, if not, process 1300 maytransition to operation 1340, by which the k index is incremented.Process 1300 may then transition to operation 1332 and continue fromthat point.

If index k is equal to the number of peaks in the Ψ-matrix, then process1300 may transition to operation 1342, by which a number of fingers of arake combiner equivalent to the number of peaks is instantiated in areceiver detector. In operation 1344, the Doppler and delay estimatesare provided to the receiver detector for coarse signal acquisition. Inoperation 1346, the adaptive channel gain estimator (embodied by CGETs,for example) is initialized with the channel gain estimates. Inoperation 1348, message bits are generated by the receiver detectorthrough coarse signal acquisition using the Doppler shift estimates, thepropagation delay estimates and the channel gain estimates.

The inventive principles described above may be embodied in a variety ofways including those set forth in the paragraphs that follow.

(1) A method of estimating channel effects on a radio signal conveyedover a communication channel, the method comprising: generatingsequential receiver signal samples by sampling a receiver signal at asampling rate equivalent to a chip rate at which chips of a known signalare timed; segmenting the receiver signal samples into receiver signalsegments; filtering the receiver signal segments by respective matchedfilters to produce respective filter response sequences, the matchedfilters being maximally responsive to respective known signal segmentssegmented from the known signal; assigning indexes to elements of thefilter response sequences to define an array thereof, the elements ofthe filter response sequences being indexed in a row order defined by asequence order of the known signal segments as distributed across thematched filters and in a column order defined by a sequence order inwhich the receiver signal samples of the receiver signal segments arefiltered; computing frequency transforms of elements of the filterresponse sequences indexed in respective columns of the array to producerespective frequency response sequences; assigning indexes to elementsof the frequency response sequences to define another array thereof, theelements of the frequency response sequences being indexed in a columnorder defined by a sequence order in which the receiver signal samplesof the receiver signal segments are filtered and in a row order definedby a sequence order of the frequency response sequences; and jointlyestimating the channel effects from characteristics of the other arrayat which at least one local maximum is located.

(2) The method (1) above, wherein the receiver signal is an informationbearing signal modulated by a pseudo noise (PN) code and having encodedthereon message bits that are timed at a message bit rate.

(3) The method (1) or (2) above, wherein the known signal segmentscomprise a number of chips of the PN code equal to a number of chipscontained in each of the message bits.

(4) The method of any one of (1) to (3) above, further comprisingaccepting, at a receiver terminating the communication channel, a valuefor a maximum expected propagation delay time.

(5) The method of any one of (1) to (4) above, wherein segmenting thereceiver signal samples comprises segmenting the receiver signal samplesto have a greater number of chips than the known signal segments, thegreater number of chips being based on the maximum expected propagationdelay time.

(6) The method of any one of (1) to (5) above, wherein assigning indexesto elements of the filter response sequences to define an array thereofcomprises indexing only those elements of the filter response sequencesthat correspond to the interval {−τ_(max), . . . , 0, . . . , τ_(max)},where τ_(max) is the maximum expected propagation delay time.

(7) The method of any one of (1) to (6) above, wherein jointlyestimating the channel effects comprises: determining a location of thelocal maximum in the other array by row and column indexes thereof; andestimating Doppler shift imparted on an information bearing signal fromthe row index of the location of the local maximum and propagation delayimparted on the information bearing signal from the column index of thelocation of the local maximum.

(8) The method of any one of (1) to (7) above, further comprisinggenerating message bits of an information-bearing signal from thereceiver signal samples using the estimated channel effects.

(9) The method of any one of (1) to (8) above, wherein jointlyestimating the channel effects comprises: determining locations of localmaxima in the other array by respective row and column indexes thereof;and estimating Doppler shift imparted on multipath copies of theinformation bearing signal from the row indexes of the local maxima andpropagation delay times imparted on the multipath copies of theinformation bearing signal from the column indexes of the local maxima.

(10) An apparatus for estimating channel effects on a radio signalconveyed over a communication channel, the apparatus comprising: memorycircuitry configured to store receiver signal samples; and processorcircuitry configured to: generate the receiver signal samples bysequentially sampling a receiver signal at a sampling rate equivalent toa chip rate at which chips of a known signal are timed; segment thereceiver signal samples into receiver signal segments; filter thereceiver signal segments by respective matched filters to producerespective filter response sequences, the matched filters beingmaximally responsive to respective known signal segments segmented fromthe known signal; assign indexes to elements of the filter responsesequences to define an array thereof, the elements of the filterresponse sequences being indexed in a row order defined by a sequenceorder of the known signal segments as distributed across the matchedfilters and in a column order defined by a sequence order in which thereceiver signal samples of the receiver signal segments are filtered;compute frequency transforms of elements of the filter responsesequences indexed in respective columns of the array to producerespective frequency response sequences; assign indexes to elements ofthe frequency response sequences to define another array thereof, theelements of the frequency response sequences being indexed in a columnorder defined by a sequence order in which the receiver signal samplesof the receiver signal segments are filtered and in a row order definedby a sequence order of the frequency response sequences; and jointlyestimate the channel effects from characteristics of the other array atwhich at least one local maximum is located.

(11) The apparatus (10) above, wherein the receiver signal is aninformation bearing signal modulated by a pseudo noise (PN) code andhaving encoded thereon message bits that are timed at a message bitrate.

(12) The apparatus of (10) or (11) above, wherein the known signalsegments comprise a number of chips of the PN code equal to a number ofchips contained in each of the message bits.

(13) The apparatus of any one of (10) to (12) above, wherein theprocessor circuitry is further configured to accept, at a receiverterminating the communication channel, a value for a maximum expectedpropagation delay time.

(14) The apparatus of any one of (10) to (13) above, wherein theprocessor circuitry is further configured to segment the receiver signalsamples to have a greater number of chips than the known signalsegments, the greater number of chips being based on the maximumexpected propagation delay time.

(15) The apparatus of any one of (10) to (14) above, wherein theprocessor circuitry is further configured to assign indexes to elementsof the filter response sequences to define an array thereof by indexingonly those elements of the filter response sequences that correspond tothe interval {−τ_(max), . . . , 0, . . . , τ_(max)}, where τ_(max) isthe maximum expected propagation delay time.

(16) The apparatus of any one of (10) to (15) above, wherein theprocessor circuitry is further configured to: determine a location ofthe local maximum in the other array by row and column indexes thereof;and estimate Doppler shift imparted on the information bearing signalfrom the row index of the location of the local maximum and propagationdelay imparted on the information bearing signal from the column indexof the location of the local maximum.

(17) An apparatus for recovering message bits representing informationin an information bearing signal conveyed over a communication channelthat imparts effects on the information bearing signal, the message bitsbeing timed at a predetermined message bit rate, the method comprising:memory circuitry configured for storing receiver signal samples; andprocessing circuitry configured to: generate the receiver signal samplesby sequentially sampling a receiver signal at a sampling rate equivalentto a chip rate at which chips of a known signal are timed; segment thereceiver signal samples into receiver signal segments; filter thereceiver signal segments by respective matched filters to producerespective filter response sequences, the matched filters beingmaximally responsive to respective known signal segments segmented fromthe known signal; assign indexes to elements of the filter responsesequences to define an array thereof, the elements of the filterresponse sequences being indexed in a row order defined by a sequenceorder of the known signal segments as distributed across the matchedfilters and in a column order defined by a sequence order in which thereceiver signal samples of the receiver signal segments are filtered;compute frequency transforms of elements of the filter responsesequences indexed in respective columns of the array to producerespective frequency response sequences; assign indexes to elements ofthe frequency response sequences to define another array thereof, theelements of the frequency response sequences being indexed in a columnorder defined by a sequence order in which the receiver signal samplesof the receiver signal segments are filtered and in a row order definedby a sequence order of the frequency response sequences; jointlyestimate the channel effects from characteristics of the other array atwhich local maxima are located; and generate the message bits from thereceiver signal samples using the estimated effects of the communicationchannel.

(18) The apparatus (17) above, wherein the processing circuitry isfurther configured to: determine locations of the local maxima in theother array by respective row and column indexes thereof; and estimateDoppler shift imparted on multipath copies of the information bearingsignal from the row indexes of the local maxima and propagation delaytimes imparted on the multipath copies of the information bearing signalfrom the column indexes of the local maxima.

(19) The apparatus of (17) or (18) above, wherein the processingcircuitry is further configured to: compensate for the propagation delaytimes by delaying the receiver samples across detector circuit paths bycorresponding delay times equal to the respective estimates thereof, thedetector circuit paths corresponding to respective propagation pathsover which the multipath copies of the information bearing signal areconveyed; refine the delay times by which the receiver samples aredelayed via a delay locked loop; and generate the message bits from thedelayed receiver samples.

(20) The apparatus of any one of (17) to (19) above, wherein theprocessing circuitry is further configured to: compensate for theDoppler shifts by modulating the receiver signal samples across thedetector circuit paths by respective frequencies corresponding to therespective estimates of the Doppler shifts; and generate the messagebits from the compensated signal samples.

(21) A method of estimating communication channel effects on aninformation bearing signal conveyed over a communication channel, theinformation bearing signal having encoded thereon message bitsrepresenting the information borne on the information bearing signalthat are timed at a message bit rate, the method comprising: generatingsequential receiver signal samples by sampling a receiver signal derivedfrom the information bearing signal at a sampling rate equivalent to achip rate at which chips of a pseudo-noise (PN) code are timed, the chiprate being a number N times the message bit rate; filtering successivesequences of the number N receiver signal samples by respective matchedfilters to produce respective filter response sequences, each of thematched filters being maximally responsive to a corresponding one ofsuccessive sequences of the number N chips of the PN code; computingfrequency transforms of elements of the filter response sequences atlike sequence locations thereof across the filter response sequences toproduce frequency response sequences, the frequency transforms beingbased on the message bit rate; assigning indexes to elements of thefrequency response sequences to define an array thereof, the elements ofthe frequency response sequences being indexed in the array in a columnorder defined by the sequence location of the elements of the filterresponse sequences for which the respective frequency transforms werecomputed and in a row order defined by a succession order of thesequences of chips of the PN code to which the respective matchedfilters that produced the respective filter response sequences aremaximally responsive; and jointly estimating the communication channeleffects from characteristics of the array at which at least one localmaximum is located.

(22) The method (21) above, wherein jointly estimating the communicationchannel effects comprises: determining a location of the local maximumin the array by row and column indexes thereof; and estimating Dopplershift imparted on the information bearing signal from the row index ofthe location of the local maximum and propagation delay imparted on theinformation bearing signal from the column index of the location of thelocal maximum.

(23) The method of (21) or (22) above, wherein jointly estimating thecommunication channel effects further comprises estimating channel gainfrom a value of the local maximum.

(24) A method of recovering message bits representing information in aninformation bearing signal conveyed over a communication channel thatimparts communication channel effects thereon, the message bits beingtimed at a predetermined message bit rate, the method comprising:generating sequential receiver signal samples by sampling a receiversignal derived from the information bearing signal at a sampling rateequivalent to a chip rate at which chips of a pseudo-noise (PN) code aretimed, the chip rate being a number N times the message bit rate;filtering successive sequences of the number N receiver signal samplesby respective matched filters to produce respective filter responsesequences, each of the matched filters being maximally responsive to acorresponding one of successive sequences of the number N chips of thePN code; computing frequency transforms of elements of the filterresponse sequences at like sequence locations thereof across the filterresponse sequences to produce frequency response sequences, thefrequency transforms being based on the message bit rate; assigningindexes to elements of the frequency response sequences to define anarray thereof, the elements of the frequency response sequences beingindexed in the array in a column order defined by the sequence locationof the elements of the filter response sequences for which therespective frequency transforms were computed and in a row order definedby a succession order of the sequences of chips of the PN code to whichthe respective matched filters that produced the respective filterresponse sequences are maximally responsive; jointly estimating thecommunication channel effects from characteristics of the array at whichlocal maxima are located; and generating the message bits from thereceiver signal samples using the estimated communication channeleffects.

(25) The method (24) above, wherein jointly estimating the communicationchannel effects comprises: determining locations of the local maxima inthe array by respective row and column indexes thereof; and estimatingDoppler shift imparted on multipath copies of the information bearingsignal from the row indexes of the local maxima and propagation delaytimes imparted on the multipath copies of the information bearing signalfrom the column indexes of the local maxima.

(26) The method of (24) or (25) above, wherein generating the messagebits from the receiver signal samples comprises: compensating for thepropagation delay times by delaying the receiver samples across detectorcircuit paths by corresponding delay times equal to the respectiveestimates thereof, the detector circuit paths corresponding torespective propagation paths over which the multipath copies of theinformation bearing signal are conveyed; refining the delay times bywhich the receiver samples are delayed via a delay locked loop; andgenerating the message bits from the delayed receiver samples.

(27) The method of any one of (24) to (26) above, wherein generating themessage bits from the receiver signal samples further comprises:compensating for the Doppler shifts by modulating the receiver signalsamples across the detector circuit paths by respective frequenciescorresponding to the respective estimates of the Doppler shifts; andgenerating the message bits from the compensated signal samples.

(28) The method of any one of (24) to (27) above, further comprising:combining the compensated receiver signal samples from each of thedetector circuit paths using maximal-ratio combining; and generating themessage bits from the combined receiver signal samples.

(29) The method of any one of (24) to (28) above, wherein jointlyestimating the communication channel effects further comprisesestimating channel gains from values of the local maxima.

(30) The method of any one of (24) to (29) above, wherein generating themessage bits from the receiver signal samples comprises: compensatingfor the channel gains using the channel gain estimates of eachpropagation path; and generating the message bits from the compensatedreceiver signal samples.

(31) An apparatus for estimating communication channel effects on aninformation bearing signal conveyed over a communication channel, theinformation bearing signal having encoded thereon message bitsrepresenting the information borne on the information bearing signalthat are timed at a message bit rate, the apparatus comprising: memorycircuitry configured to store receiver signal samples; and processorcircuitry configured to: generate the receiver signal samples bysampling a receiver signal derived from the information bearing signalat a sampling rate equivalent to a chip rate at which chips of apseudo-noise (PN) code are timed, the chip rate being a number N timesthe message bit rate; filter successive sequences of the number Nreceiver signal samples by respective matched filters to producerespective filter response sequences, each of the matched filters beingmaximally responsive to a corresponding one of successive sequences ofthe number N chips of the PN code; compute frequency transforms ofelements of the filter response sequences at like sequence locationsthereof across the filter response sequences to produce frequencyresponse sequences, the frequency transforms being based on the messagebit rate; assign indexes to elements of the frequency response sequencesto define an array thereof, the elements of the frequency responsesequences being indexed in the array in a column order defined by thesequence location of the elements of the filter response sequences forwhich the respective frequency transforms were computed and in a roworder defined by a succession order of the sequences of chips of the PNcode to which the respective matched filters that produced therespective filter response sequences are maximally responsive; andjointly estimate the communication channel effects from characteristicsof the array at which at least one local maximum is located.

(32) The apparatus (31) above, wherein the processor circuitry isfurther configured to: determine a location of the local maximum in thearray by row and column indexes thereof; and estimate Doppler shiftimparted on the information bearing signal from the row index of thelocation of the local maximum and propagation delay imparted on theinformation bearing signal from the column index of the location of thelocal maximum.

(33) The apparatus of (31) or (32) above, wherein the processorcircuitry is further configured to estimate channel gain from a value ofthe local maximum.

(34) An apparatus for recovering message bits representing informationin an information bearing signal conveyed over a communication channelthat imparts effects on the information bearing signal, the message bitsbeing timed at a predetermined message bit rate, the method comprising:memory circuitry configured for storing receiver signal samples; andprocessing circuitry configured to: generate the receiver signal samplesby sampling a receiver signal derived from the information bearingsignal at a sampling rate equivalent to a chip rate at which chips of apseudo-noise (PN) code are timed, the chip rate being a number N timesthe message bit rate; filter successive sequences of the number Nreceiver signal samples by respective matched filters to producerespective filter response sequences, each of the matched filters beingmaximally responsive to a corresponding one of successive sequences ofthe number N chips of the PN code; compute frequency transforms ofelements of the filter response sequences at like sequence locationsthereof across the filter response sequences to produce frequencyresponse sequences, the frequency transforms being based on the messagebit rate; assign indexes to elements of the frequency response sequencesto define an array thereof, the elements of the frequency responsesequences being indexed in the array in a column order defined by thesequence location of the elements of the filter response sequences forwhich the respective frequency transforms were computed and in a roworder defined by a succession order of the sequences of chips of the PNcode to which the respective matched filters that produced therespective filter response sequences are maximally responsive; jointlyestimate the communication channel effects from characteristics of thearray at which local maxima are located; and generate the message bitsfrom the receiver signal samples using the estimated effects of thecommunication channel.

(35) The apparatus (34) above, wherein the processing circuitry isfurther configured to: determine locations of the local maxima in thearray by respective row and column indexes thereof; and estimate Dopplershift imparted on multipath copies of the information bearing signalfrom the row indexes of the local maxima and propagation delay timesimparted on the multipath copies of the information bearing signal fromthe column indexes of the local maxima.

(36) The apparatus (34) or (35) above, wherein the processing circuitryis further configured to: compensate for the propagation delay times bydelaying the receiver samples across detector circuit paths bycorresponding delay times equal to the respective estimates thereof, thedetector circuit paths corresponding to respective propagation pathsover which the multipath copies of the information bearing signal areconveyed; refine the delay times by which the receiver samples aredelayed via a delay locked loop; and generate the message bits from thedelayed receiver samples.

(37) The apparatus of any one of (34) to (36) above, wherein theprocessing circuitry is further configured to: compensate for theDoppler shifts by modulating the receiver signal samples across thedetector circuit paths by respective frequencies corresponding to therespective estimates of the Doppler shifts; and generate the messagebits from the compensated signal samples.

(38) The apparatus of any one of (34) to (37) above, wherein theprocessing circuitry is further configured to: combine the compensatedreceiver signal samples from each of the detector circuit paths usingmaximal-ratio combining; and generate the message bits from the combinedreceiver signal samples.

(39) The apparatus of any one of (34) to (38) above, wherein theprocessing circuitry is further configured to estimate channel gainsfrom values of the local maxima.

(40) The apparatus of any one of (34) to (39) above, wherein theprocessing circuitry is further configured to: compensate for thechannel gains using the channel gain estimates of each propagation path;and generate the message bits from the compensated receiver signalsamples.

The terminology used herein is for the purpose of describing particularembodiments only and is not intended to be limiting of the principlesdescribed herein. As used herein, the singular forms “a”, “an” and “the”are intended to include the plural forms as well, unless the contextclearly indicates otherwise. It will be further understood that theterms “comprises” and/or “comprising,” when used in this specification,specify the presence of stated features, integers, steps, operations,elements, and/or components, but do not preclude the presence oraddition of one or more features, integers, steps, operations, elements,components, and/or groups thereof.

The corresponding structures, materials, acts, and equivalents of allmeans or step plus function elements in the claims below are intended toinclude any structure, material, or act for performing the function incombination with other claimed elements as specifically claimed. Thedescription in this disclosure has been presented for purposes ofillustration and description, but is not intended to be exhaustive orlimited to the embodiments disclosed. The embodiments were chosen anddescribed in order to best explain the principles of the underlyingconcept and the practical application, and to enable others of ordinaryskill in the art to understand the principles for various embodimentswith various modifications as are suited to the particular usecontemplated.

The descriptions above are intended to illustrate possibleimplementations of the present inventive concept and are notrestrictive. Many variations, modifications and alternatives will becomeapparent to the skilled artisan upon review of this disclosure. Forexample, components equivalent to those shown and described may besubstituted therefore, elements and methods individually described maybe combined, and elements described as discrete may be distributedacross many components. The scope of the disclosure should therefore bedetermined not with reference to the description above, but withreference to the appended claims, along with their full range ofequivalents.

1. A method of estimating channel effects on a radio signal conveyedover a communication channel, the method comprising: generatingsequential receiver signal samples by sampling a receiver signal at asampling rate equivalent to a chip rate at which chips of a known signalare timed; segmenting the receiver signal samples into receiver signalsegments; filtering the receiver signal segments by respective matchedfilters to produce respective filter response sequences, the matchedfilters being maximally responsive to respective known signal segmentssegmented from the known signal; assigning indexes to elements of thefilter response sequences to define an array thereof, the elements ofthe filter response sequences being indexed in a first row order definedby a first sequence order of the known signal segments as distributedacross the matched filters and in a first column order defined by asecond sequence order in which the receiver signal samples of thereceiver signal segments are filtered; computing frequency transforms ofelements of the filter response sequences indexed in respective columnsof the array to produce respective frequency response sequences;assigning indexes to elements of the frequency response sequences todefine another array thereof, the elements of the frequency responsesequences being indexed in a second column order defined by a thirdsequence order in which the receiver signal samples of the receiversignal segments are filtered and in a second row order defined by afourth sequence order of the frequency response sequences; and jointlyestimating the channel effects from characteristics of the other arrayat which at least one local maximum is located.
 2. The method of claim1, wherein the receiver signal is an information bearing signalmodulated by a pseudo noise (PN) code and having encoded thereon messagebits that are timed at a message bit rate.
 3. The method of claim 2,wherein the known signal segments comprise a number of chips of the PNcode equal to a number of chips contained in each of the message bits.4. The method of claim 1, further comprising accepting, at a receiverterminating the communication channel, a value for a maximum expectedpropagation delay time.
 5. The method of claim 4, wherein the segmentingof the receiver signal samples comprises segmenting the receiver signalsamples to have a greater number of chips than the known signalsegments, the greater number of chips being based on the maximumexpected propagation delay time.
 6. The method of claim 5, wherein theassigning of indexes to elements of the filter response sequences todefine the array thereof comprises indexing only those elements of thefilter response sequences that correspond to the interval {−τ_(max), . .. , 0, . . . , τ_(max)}, where τ_(max) is the maximum expectedpropagation delay time.
 7. The method of claim 1, wherein the jointlyestimating of the channel effects comprises: determining a location ofthe local maximum in the other array by a row index and a column indexthereof; and estimating Doppler shift imparted on an information bearingsignal from the row index of the location of the local maximum andpropagation delay imparted on the information bearing signal from thecolumn index of the location of the local maximum.
 8. The method ofclaim 1, further comprising generating message bits of an informationbearing signal from the receiver signal samples using the jointlyestimated channel effects.
 9. The method of claim 8, wherein the jointlyestimating of the channel effects comprises: determining locations oflocal maxima in the other array by respective row and column indexesthereof; and estimating Doppler shift imparted on multipath copies ofthe information bearing signal from the row indexes of the local maximaand propagation delay times imparted on the multipath copies of theinformation bearing signal from the column indexes of the local maxima.10. An apparatus for estimating channel effects on a radio signalconveyed over a communication channel, the apparatus comprising: memorycircuitry configured to store receiver signal samples; and processorcircuitry configured to: generate the receiver signal samples bysequentially sampling a receiver signal at a sampling rate equivalent toa chip rate at which chips of a known signal are timed; segment thereceiver signal samples into receiver signal segments; filter thereceiver signal segments by respective matched filters to producerespective filter response sequences, the matched filters beingmaximally responsive to respective known signal segments segmented fromthe known signal; assign indexes to elements of the filter responsesequences to define an array thereof, the elements of the filterresponse sequences being indexed in a first row order defined by a firstsequence order of the known signal segments as distributed across thematched filters and in a first column order defined by a second sequenceorder in which the receiver signal samples of the receiver signalsegments are filtered; compute frequency transforms of elements of thefilter response sequences indexed in respective columns of the array toproduce respective frequency response sequences; assign indexes toelements of the frequency response sequences to define another arraythereof, the elements of the frequency response sequences being indexedin a second column order defined by a third sequence order in which thereceiver signal samples of the receiver signal segments are filtered andin a second row order defined by a fourth sequence order of thefrequency response sequences; and jointly estimate the channel effectsfrom characteristics of the other array at which at least one localmaximum is located.
 11. The apparatus of claim 10, wherein the receiversignal is an information bearing signal modulated by a pseudo noise (PN)code and having encoded thereon message bits that are timed at a messagebit rate.
 12. The apparatus of claim 11, wherein the known signalsegments comprise a number of chips of the PN code equal to a number ofchips contained in each of the message bits.
 13. The apparatus of claim10, wherein the processor circuitry is further configured to accept, ata receiver comprising the processor circuitry and terminating thecommunication channel, a value for a maximum expected propagation delaytime.
 14. The apparatus of claim 13, wherein the processor circuitry isfurther configured to segment the receiver signal samples to have agreater number of chips than the known signal segments, the greaternumber of chips being based on the maximum expected propagation delaytime.
 15. The apparatus of claim 14, wherein the processor circuitry isfurther configured to assign indexes to elements of the filter responsesequences to define the array thereof by indexing only those elements ofthe filter response sequences that correspond to the interval {−τ_(max),. . . , 0, . . . , τ_(max)}, where τ_(max) is the maximum expectedpropagation delay time.
 16. The apparatus of claim 10, wherein theprocessor circuitry is further configured to: determine a location ofthe local maximum in the other array by a row index and a column indexthereof; and estimate Doppler shift imparted on the information bearingsignal from the row index of the location of the local maximum andpropagation delay imparted on the information bearing signal from thecolumn index of the location of the local maximum.
 17. An apparatus forrecovering message bits representing information in an informationbearing signal conveyed over a communication channel that impartseffects on the information bearing signal, the message bits being timedat a predetermined message bit rate, the apparatus comprising: memorycircuitry configured for storing receiver signal samples; and processingcircuitry configured to: generate the receiver signal samples bysequentially sampling a receiver signal at a sampling rate equivalent toa chip rate at which chips of a known signal are timed; segment thereceiver signal samples into receiver signal segments; filter thereceiver signal segments by respective matched filters to producerespective filter response sequences, the matched filters beingmaximally responsive to respective known signal segments segmented fromthe known signal; assign indexes to elements of the filter responsesequences to define an array thereof, the elements of the filterresponse sequences being indexed in a first row order defined by a firstsequence order of the known signal segments as distributed across thematched filters and in a first column order defined by a second sequenceorder in which the receiver signal samples of the receiver signalsegments are filtered; compute frequency transforms of elements of thefilter response sequences indexed in respective columns of the array toproduce respective frequency response sequences; assign indexes toelements of the frequency response sequences to define another arraythereof, the elements of the frequency response sequences being indexedin a second column order defined by a third sequence order in which thereceiver signal samples of the receiver signal segments are filtered andin a second row order defined by a fourth sequence order of thefrequency response sequences; jointly estimate the channel effects fromcharacteristics of the other array at which local maxima are located;and generate the message bits from the receiver signal samples using theestimated effects of the communication channel.
 18. The apparatus ofclaim 17, wherein the processing circuitry is further configured to:determine locations of the local maxima in the other array by respectiverow and column indexes thereof; and estimate Doppler shift imparted onmultipath copies of the information bearing signal from the row indexesof the local maxima and propagation delay times imparted on themultipath copies of the information bearing signal from the columnindexes of the local maxima.
 19. The apparatus of claim 18, wherein theprocessing circuitry is further configured to: compensate for thepropagation delay times by delaying the receiver samples across detectorcircuit paths by corresponding delay times equal to the respectiveestimates thereof, the detector circuit paths corresponding torespective propagation paths over which the multipath copies of theinformation bearing signal are conveyed; refine the delay times by whichthe receiver samples are delayed via a delay locked loop; and generatethe message bits from the delayed receiver samples.
 20. The apparatus ofclaim 19, wherein the processing circuitry is further configured to:compensate for the Doppler shifts by modulating the receiver signalsamples across the detector circuit paths by respective frequenciescorresponding to the respective estimates of the Doppler shifts; andgenerate the message bits from the compensated signal samples.